Method and system for calibration in an fm transceiver system

ABSTRACT

Aspects of a method and system for calibration in an FM transceiver system may include: in an integrated FM system comprising an FM radio transmitter, an FM radio receiver and a common local oscillator, generating via the common local oscillator, one or more RF carrier signals and corresponding phase-shifted versions of the generated one or more RF carrier signals. The FM radio transmitter and/or the FM radio receiver may be calibrated based on an RF calibration signal generated from the one or more RF carrier signals and/or the corresponding phase-shifted versions of the generated one or more RF carrier signals. A phase between the one or more RF carrier signals and the corresponding phase-shifted versions of the generated one or more RF carrier signals may be adjusted based on the generated RF calibration signal. An in-phase baseband signal component associated with the generated RF calibration signal may be zeroed.

CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY REFERENCE

This application makes reference to, claims priority to, and claims the benefit of U.S. Provisional Application Ser. No. 60/895,665, filed on Mar. 19, 2007.

The above referenced application is hereby incorporated herein by reference in its entirety.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to signal processing for communication systems. More specifically, certain embodiments of the invention relate to a method and system for calibration in an FM transceiver system.

BACKGROUND OF THE INVENTION

Electronic communication has become prolific over the last decade. While electronic communication was initially limited to the desktop, recent trends have been to make communications, media content and the Internet available anytime, anywhere and, increasingly, on any device. Already now, it is quite common to find mobile devices such as cellular phones or Personal Digital Assistants (PDAs) that incorporate a large range of communication technologies and associated software. For example, fully-featured web-browsers, email clients, MP3 players, instant messenger software, and Voice-over-IP may all be found on some recent devices.

In this same spirit of the ‘anytime, anywhere’ paradigm, there is a drive towards making portable devices ever more capable and smaller, while making stored content available on a variety of displays and user interfaces. For example, many portable media devices may be enabled to provide a video output signal to a computer monitor or a television to allow display of, for example, digital photographs. For audio content, one possible output format may be a low-power FM transmission signal. Recent changes, for example, in European regulation by CEPT/ETSI to the category of Short Range Devices (SDR) may now permit the use of very low power FM transmitters to transmit in the FM radio broadcast spectrum at powers of around 50 nW.

Further limitations and disadvantages of conventional and traditional approaches will become apparent to one of skill in the art, through comparison of such systems with some aspects of the present invention as set forth in the remainder of the present application with reference to the drawings.

BRIEF SUMMARY OF THE INVENTION

A method and/or system for calibration in an FM transceiver system, substantially as shown in and/or described in connection with at least one of the figures, as set forth more completely in the claims.

These and other advantages, aspects and novel features of the present invention, as well as details of an illustrated embodiment thereof, will be more fully understood from the following description and drawings.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1 is a block diagram illustrating an exemplary FM transceiver system, in accordance with an embodiment of the invention.

FIG. 2A is a block diagram illustrating an exemplary FM calibration system, in accordance with an embodiment of the invention.

FIG. 2B is a block diagram illustrating an exemplary FM calibration system with a single programmable phase shift, in accordance with an embodiment of the invention.

FIG. 3 is a flow chart illustrating an exemplary sequential I/Q mismatch calibration protocol, in accordance with an embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

Certain embodiments of the invention may be found in a method and system for calibration in an FM transceiver system. Aspects of the method and system for calibration in an FM transceiver system may comprise: in an integrated FM system comprising an FM radio transmitter, an FM radio receiver and a common local oscillator, generating via the common local oscillator, one or more RF carrier signals and corresponding phase-shifted versions of the generated one or more RF carrier signals. The FM radio transmitter and/or the FM radio receiver may be calibrated based on an RF calibration signal generated from the one or more RF carrier signals and/or the corresponding phase-shifted versions of the generated one or more RF carrier signals.

A phase between the one or more RF carrier signals and the corresponding phase-shifted versions of the generated one or more RF carrier signals may be adjusted based on the generated RF calibration signal. An in-phase baseband signal component associated with the generated RF calibration signal may be zeroed. A signal power of an in-phase signal component after demodulating the generated RF calibration signal may be measured. A quadrature baseband signal component associated with the generated RF calibration signal may be zeroed. A signal power of a quadrature signal component after demodulating the generated RF calibration signal may be measured. Based on the generated RF calibration signal, a tuning range and/or gain stages of the FM radio receiver and/or the FM radio transmitter may be adjusted and/or calibrated.

FIG. 1 is a block diagram illustrating an exemplary FM transceiver system, in accordance with an embodiment of the invention. Referring to FIG. 1, there is shown an FM transceiver system 100 comprising an antenna 102, a coupler 104, an FM receiver 150, an FM transmitter 180, a device control 106, a transmitter-receiver (TX/RX) matching block 110, a local oscillator 108 and a switch 112.

The FM transceiver system 100 may comprise suitable logic, circuitry and/or code that may be enabled to transmit and receive FM signals simultaneously on different frequencies and/or in an alternating fashion on the same frequency. The FM transmitter 180 may comprise suitable logic, circuitry and/or code to enable generation of a transmit signal that may be communicated to the coupler 104. The FM receiver 150 may comprise suitable logic, circuitry and/or logic that may enable reception and/or processing of FM signals, fed to it from the coupler 104. The antenna 102 may be a shared antenna for a transmit signal path and a receive signal path. The transmit signal path from the FM transmitter 180 and the receive signal path to the FM receiver 150 may be coupled to the antenna 102 at the coupler 104 that may comprise suitable logic, circuitry and/or code to join the receive signal path and the transmit signal path, in order to communicatively couple a common signal path to antenna 102. A device control block 106 may comprise suitable logic, circuitry and/or code to enable controlling the FM transmitter 180 and the FM receiver 150, as well as the TX/RX matching block 110 and the local oscillator 108. The control block 106 may control, for example, a gain and/or a demodulation frequency in the FM receiver 150 via the local oscillator 108 and, for example, a transmit power and transmit frequency for the FM transmitter 180. The functionality of the device control block 106 may not be limited to the functionality described above. The local oscillator 108 may comprise suitable logic, circuitry and/or code that may be enabled to generate oscillating signals that may be utilized, for example, in modulation and demodulation of radio-frequency (RF) signals in the FM receiver 150 and the FM transmitter 180. The local oscillator 108 may drive the FM receiver 150 and/or the FM transmitter 180 with one or more carrier signals at different frequencies.

In some instances, for example when the FM transceiver system 100 may be initialized, it may be desirable to perform self-tests and/or calibration of the FM receiver 150 and/or the FM transmitter 180. Since an FM receiver 150 and an FM transmitter 180 may be integrated in the FM transceiver system 100, a transmit RF signal that may be generated in the FM transmitter 180 may be communicatively coupled to the FM receiver 150 for self-tests and/or calibration purposes via the switch 112 and the TX/RX matching block 110. By enabling the switch 112, the transmit signal from the FM transmitter 180 may be communicated to the FM receiver 150 and may bypass the antenna 102 and the coupler 104. The TX/RX matching block 110 may comprise suitable logic, circuitry and/or code that may be enabled to provide an appropriate RF coupling between the FM transmitter 180 and the FM receiver 150. The TX/RX matching block 110 may, for example, match the output impedance of the FM transmitter 180 and/or the input impedance of the FM receiver 150. In another embodiment of the invention, a TX/RX matching block 110 may not be required.

In various other embodiments of the invention, the FM receiver 150 and the FM transmitter 180 may use separate antennas. In various other embodiments of the invention, the FM transceiver system 100 may be a stand-alone system or may form part of a device, for example, a personal audio player or a cellular mobile phone. In one embodiment of the invention, the FM transceiver system 100 may be integrated in a single chip, comprising the FM transmitter 180 and the FM receiver 150. The invention may not be limited to the examples given above.

FIG. 2A is a block diagram illustrating an exemplary FM calibration system, in accordance with an embodiment of the invention. Referring to FIG. 2A, there is shown a device control block 206, a local oscillator 208, a switch 212, a transmitter (TX) programmable phase shift 202, a receiver (RX) programmable phase shift 204, a TX RF modulator 280, an RX RF demodulator 250, an I-channel received signal strength indicator (RSSI) block 214 and a Q-channel RSSI block 210. The TX RF modulator 280 may comprise multipliers 282 and 286, and adder 284. The RX RF demodulator may comprise multipliers 252 and 256, amplifiers 258 and 260, and low-pass filters (LPF) 262 and 264. There is also shown a transmit signal s(t), a transmit in-phase (I) baseband signal s_(I)(t), a transmit quadrature (Q) baseband signal s_(Q)(t), a receive in-phase baseband signal r_(I)(t), a receive quadrature baseband signal r_(Q)(t), an unfiltered demodulated I-channel signal u_(I)(t), an unfiltered demodulated Q-channel signal u_(Q)(t), and local oscillator output signals f_(tx) and f_(rx).

The diagram in FIG. 2A illustrates an exemplary system configuration that may be used to calibrate the I-channel and the Q-channel, in order to reduce I/Q channel mismatch. The I/Q channel mismatch may occur when the carrier for the I-channel and the carrier for the Q-channel may be misaligned in phase at the transmitter and/or the receiver. The transmit signal s(t) may be generated in the TX RF modulator 280 by modulating s_(I)(t) onto the I-channel carrier in multiplier 282 and modulating S_(Q)(t) onto the Q-channel carrier in multiplier 286, and summing the modulated signal components in adder 284. The multipliers 282 and 286 may comprise suitable logic, circuitry and/or code that may be enabled to multiply a carrier signal with a baseband signal to obtain frequency translation, for example, for modulation purposes. The adder 284 may comprise suitable logic, circuitry and/or code that may be enabled to add two time-domain signals and generate an output signal that may be proportional to the sum of the two input signals. The I-channel carrier cos(w_(c)t)=f_(tx) may be fed to the multiplier 282 from the local oscillator 208. The local oscillator 208 may comprise suitable logic, circuitry and/or code that may be enabled to generate variable frequency sinusoidal output signals, for example f_(rx) and f_(tx), that may be used as radio-frequency carriers.

In one embodiment of the invention, the local oscillator may generate multiple output frequencies that may be different and programmable for each individual output, providing different carrier frequencies f_(rx) and f_(tx) to the RX RF demodulator 250 and the TX RF modulator 280, respectively. Notwithstanding, for the embodiment of the invention described in FIG. 2A, f_(rx)=f_(tx)=cos(w_(c)t). The Q-channel carrier −sin(w_(c)t+e_(t)) may be fed to the multiplier 286 from the TX programmable phase shift 202. The TX programmable phase shift 202 may comprise suitable logic, circuitry and/or code that may be enabled to phase-shift the signal at its input and generate an output signal that may be a phase-shifted version of the input signal. Generally, it may be desirable that the Q-channel carrier may be 90 degrees phase-shifted from the I-channel carrier. Hence, the TX programmable phase shift 202 may generally be set to a phase shift of +90 degrees (π/2) since the desired Q-channel carrier may be obtained from the I-channel carrier cos(w_(c)t), as shown in the following relationship:

${- {\sin \left( {w_{c}t} \right)}} = {\cos\left( {{w_{c}t} + \frac{\pi}{2}} \right)}$

where w_(c)=2πf_(c) may be the angular frequency of the sinusoidal carrier signal. Ideally, a desirable transmit signal S_(d)(t) may hence be given by the following relationship:

S _(d)(t)=s _(I)(t)cos(w _(c) t)+s _(Q)(t)sin(w _(c) t)

In practice, however, the TX programmable phase shift 202 may diverge slightly from 90 degree phase shift, for example due to manufacturing tolerances of its components, changes in the operating environment etc. Hence, the actual transmit signal s(t) may comprise a phase error term e_(t), as shown in the following relationship:

s(t)=sI₍ t)cos(w _(c) t)−s _(Q)(t)sin(w _(c) t+e _(t))

In instances where a phase error e_(t) may be present, the separation of the I-channel and the Q-channel at the RX RF demodulator may be imperfect and some interference between the Q-channel and the I-channel may result.

The transmit signal s(t) that may be generated by the TX RF modulator 280 may be coupled to the RX RF demodulator 250 via switch 212. As illustrated in FIG. 2A, the switch may couple the transmit signal to the TX/RX antennas or to the RX RF demodulator 250. The switch 212 may be substantially similar to the switch 112. For self-tests, calibration and/or I/Q mismatch adjustments, the switch 212 may be in the position depicted, coupling the TX RF modulator 280 to the RX RF demodulator 250.

To demodulate the transmit signal s(t) at the RX RF demodulator 250, the RX RF demodulator 250 may utilize the same local oscillator block 208 that may be used for the TX RF modulator 280. In another embodiment of the invention, the local oscillator may comprise additional logic, circuitry and/or code to generate a different carrier frequency for the RX RF demodulator 250 that may enable simultaneous transmit and receive functionality by the transceiver system on a plurality of frequencies. In another embodiment of the invention, the RX programmable phase shift 204 and the TX programmable phase shift 202 may be replaced by a common programmable phase shift block.

At the RX RF demodulator 250, the transmit signal s(t) may be demodulated to obtain the baseband received signals r_(I)(t) and r_(Q)(t) for the I-channel and the Q-channel, respectively. Ideally, in some instances, the demodulated signals may be r_(I)(t)=s_(I)(t) and r_(Q)(t)=s_(Q)(t). The I-channel demodulation may be given by the following relationship:

$\begin{matrix} \begin{matrix} {{u_{I}(t)} = {2{\cos \left( {w_{c}t} \right)}{s(t)}}} \\ {= {2\; {{\cos \left( {w_{c}t} \right)}\left\lbrack {{{s_{I}(t)}{\cos \left( {w_{c}t} \right)}} - {{s_{Q}(t)}{\sin \left( {{w_{c}t} + e_{t}} \right)}}} \right\rbrack}}} \\ {= {{{s_{I}(t)}\left\lbrack {1 + {\cos \left( {2w_{c}t} \right)}} \right\rbrack} - {{s_{Q}(t)}\left\lbrack {{\sin \left( e_{t} \right)} + {\sin \left( {{2w_{c}t} + e_{t}} \right)}} \right\rbrack}}} \end{matrix} & (1) \end{matrix}$

where the unfiltered demodulated I-channel signal u_(I)(t) may be the signal at the input of the LPF 262. In equation (1) above, the multiplication with the I-channel carrier may be performed in the multiplier 252 that may be substantially similar to the multiplier 282. The multiplication factor 2 may be introduced with the amplifier 258, for example. The amplifier 258 may comprise suitable logic, circuitry and/or code that may be enabled to amplify the signal at its input. The unfiltered demodulated I-channel u_(I)(t) may comprise signal components at baseband, angular frequency 2 w _(c)t and e_(t), as may be seen from equation (1) above. The transmitter phase error e_(t) may be comparatively small and hence, the frequency of sin(e_(t)) may be close to baseband. The unfiltered demodulated signal u_(I)(t) may be fed to the LPF 262 in order to reject the high-frequency signal components at 2 w _(c)t, as shown in the following relationship:

r _(I)(t)=LPF[u _(I)(t)]=s _(I)(t)−s _(Q)(t)sin(e _(t))  (2)

The signal r_(I)(t) may be communicatively coupled from the LPF 262 to the I-channel RSSI block 214, which may comprise suitable logic, circuitry and/or code that may be enabled to measure the signal power at its input. The output of the I-channel RSSI block 214 may be fed to the device control 206 for further processing. The device control 206 may be substantially similar to the device control 106.

Hence, it may be observed from equation (2) that an error e_(t) in the phase difference between the I-channel carrier and the Q-channel carrier at the transmitter may introduce an error component that may be the quadrature baseband signal s_(Q)(t) modulated onto a carrier due to the error. If the error e_(t) is zero, then the signal component due to s_(Q)(t) may be eliminated. By zeroing the baseband in-phase signal s_(I)(t)=0, the remaining signal component may be due to s_(Q)(t)sin(e_(t)). Hence, by setting s_(I)(t)=0 and measuring the signal power in the I-channel RSSI block 214, the signal power of the undesirable signal component may be approximately measured and reported to the device control block 206.

The TX programmable phase shift 202 may be controlled, for example, by the device control block 206. The device control block 206 may adjust the TX programmable phase shift 202 in order to minimize the power reported to the device control block 206 from the I-channel RSSI block 214. A desirable setting for the TX programmable phase shift 202 may be corresponding to a minimum measured power that may be reported by the I-channel RSSI block 214. This setting may correspond to e_(t) approximately equal to zero, as illustrated in equation (2). These operations, in accordance with an embodiment of the invention, may hence be used to calibrate the transmitter/modulator phase shift in the TX programmable phase shift 202 and to match the I-channel and the Q-channel for the transmitter. In most instances, the baseband signal s_(Q)(t) may be chosen a constant power signal, for example a pilot tone, for I/Q mismatch calibration purposes.

Similar to the I/Q mismatch adjustment of the TX RF modulator 280 via the TX programmable phase shift block 202, the RX programmable phase shift block 204 may be adjusted via the device control 206 to reduce the I/Q mismatch in the RX RF demodulator 250. While the TX programmable phase shift 202 may be adjusted based on the signal measured by the I-channel RSSI block 214, the RX programmable phase shift 204 may be adjusted by measuring a signal power in the Q-channel via the Q-channel RSSI block 210. The transmit signal s(t) may be communicatively coupled to the multiplier 256.

The multiplier 256 may multiply the transmit signal s(t) with a shifted carrier frequency and may demodulate the quadrature component of s(t) to recover the baseband quadrature signal S_(Q)(t). The shifted carrier may be given by −sin(w_(c)t+e_(r)), similar to the shifted carrier use in multiplier 286. In various embodiments of the invention, it may be desirable to have the error e_(r) as close to zero as possible. The multiplier 256 and the amplifier 260 may be substantially similar to the multiplier 258 and the LPF 262. After multiplication and amplification of the transmit signal s(t) in the multiplier 256 and the amplifier 260, the signal u_(Q)(t) may be given by the following relationship:

$\begin{matrix} {{u_{Q}(t)} = {{- 2}\; {\sin \left( {{w_{c}t} + e_{r}} \right)}{s(t)}}} \\ {= {{- 2}{{\sin \left( {{w_{c}t} + e_{r}} \right)}\left\lbrack {{{s_{I}(t)}{\cos \left( {w_{c}t} \right)}} - {{s_{Q}(t)}{\sin \left( {{w_{c}t} + e_{t}} \right)}}} \right\rbrack}}} \\ {= {{{s_{Q}(t)}\left\lbrack {{\cos \left( {e_{t} + r_{r}} \right)} - {\cos \left( {{2w_{c}t} + e_{t} + e_{r}} \right)}} \right\rbrack} -}} \\ {{{s_{I}(t)}\left\lbrack {{\sin \left( {{2w_{c}t} + e_{r}} \right)} + {\sin \left( e_{r} \right)}} \right\rbrack}} \end{matrix}$

where e_(r) may be the phase error that may be introduced by the RX programmable phase shift 204 and e_(t) may be the phase error that may be introduced by the TX programmable phase shift 202. The signal u_(Q)(t) may be fed to the LPF 264, which may reduce the high-frequency signal components, substantially similar to LPF 262. After filtering in the LPF 264, the signal r_(Q)(t) may be obtained, as given in the following relationship:

r _(Q)(t)=LPF[u _(Q)(t)]=s _(Q)(t)cos(e _(t) +e _(r))−s _(I)(t)sin(e _(r))  (3)

From equation (3), it may be observed that r_(Q)(t) may comprise a component due to s_(Q)(t) that may depend on the phase error e_(t) at the TX programmable phase shift 202 and e_(r), the phase error at the RX programmable phase shift 204. Furthermore, r_(Q)(t) may comprise a signal component due to s_(I)(t). From equation (3), it may be observed that setting s_(Q)(t)=0 may render r_(Q)(t) a function of e_(r) and s_(I)(t). Hence, setting s_(Q)(t)=0 may permit the Q-channel RSSI block 210 to measure the approximate power of the signal component due to s_(I)(t)sin(e_(r)). By adjusting the phase in the RX programmable phase shift 204, the phase error e_(r) may be reduced and hence a desirable phase setting for the RX programmable phase shift 204 may be reached when the signal power measured in the Q-channel RSSI block 210 may be minimized. Similarly to the adjustment of the TX programmable phase shift 202, the device control 206 may adjust the phase of the RX programmable phase shift 204 in order to achieve a desirable signal power level as may be reported via the Q-channel RSSI block 210.

FIG. 2B is a block diagram illustrating an exemplary FM calibration system with a single programmable phase shift, in accordance with an embodiment of the invention. Referring to FIG. 2B, there is shown a device control block 206 b, a local oscillator 208 b, a switch 212 b, a programmable phase shift 202 b, a TX RF modulator 280 b, an RX RF demodulator 250 b, and an I-channel received signal strength indicator (RSSI) block 214 b. The TX RF modulator 280 b may comprise multipliers 282 b and 286 b and adder 284 b. The RX RF demodulator 250 b may comprise multipliers 252 b and 256 b, amplifiers 258 b and 260 b, and low-pass filters (LPF) 262 b and 264 b. There is also shown a transmit signal s(t), a transmit in-phase baseband signal s_(I)(t), a transmit quadrature baseband signal s_(Q)(t), a receive in-phase baseband signal r_(I)(t), a receive quadrature baseband signal r_(Q)(t), an unfiltered demodulated I-channel signal u_(I)(t), and an unfiltered demodulated Q-channel signal u_(Q)(t).

The numbered functional blocks and/or components in FIG. 2B with a post-fix ‘b’ may be substantially similar to the corresponding functional block in FIG. 2A without a post-fix ‘b’. For example, amplifier 260 b in FIG. 2B may be substantially similar to amplifier 260.

In various embodiments of the invention, an FM transmitter, for example the FM transmitter 180 and an FM receiver, for example FM receiver 150, may be operated in a Time Division Duplex (TDD) mode, that is, the FM transceiver system 100 may operate in receiver mode or in transmitter mode at any one given time, but not in receiver mode and transmitter mode simultaneously. In this case, for example, it may be possible to operate the TX RF modulator 280 b and the RX RF demodulator with a single carrier frequency output signal from the local oscillator 208 b. In these instances, a single programmable phase shift 202 b may be used to generate the quadrature carrier for both the RX RF demodulator 250 b and the TX RF modulator 280 b. Since there may be one programmable phase shift 202 b, a single calibration substantially similar to the calibration of the TX programmable phase shift 202 as described in FIG. 2A may be performed and it may suffice to measure the signal power of r_(I)(t) in the I-channel RSSI block 214 b.

Since an FM transceiver system in accordance with an embodiment of the invention may comprise both a transmitter and receiver, other self-tests and/or calibrations may be enabled. For example, it may be possible to test and/or calibrate the tuning range of, for example, the receiver RF demodulator 250. By generating a transmit signal s(t) at the transmitter RF modulator 280 over a certain range of frequencies, the tuning range of the RX RF demodulator 250 may be verified and/or calibrated. For example, a transmit signal s(t) may be modulated with the in-phase baseband signals s_(I)(t) and S_(Q)(t) chosen to be constant power pilot tones. Using the I-channel RSSI block 214 and the Q-channel RSSI block 210, the power in the received signal may be measured and the receive power may, for example, be adjusted by varying the phase slightly of the RX programmable phase shift 204, in order to achieve a desirable power output.

In another embodiment of the invention, a pilot signal may be comprised in s(t) that may be used to calibrate various gain stages in the FM receiver 150. For example, by generating a signal s(t) with, for example, a constant power, the amplifiers 258 and 260, for example, may be calibrated to provide desirable gain as a function of the transmit signal s(t).

FIG. 3 is a flow chart illustrating an exemplary sequential I/Q mismatch calibration protocol, in accordance with an embodiment of the invention. The calibration protocol may be started in step 302. The collection of steps 380 may represent the steps that may enable the phase of the RF modulator to be calibrated, substantially similarly to the adjustment of the TX programmable phase shift 202 as illustrated in FIG. 2A. The collection of steps 350 may represent the steps that may enable the phase of the RF demodulator to be calibrated, substantially similarly to the adjustment of the RX programmable phase shift 204 as illustrated in FIG. 2A. In various embodiment of the invention, the steps 350 and 380 may be performed in any desired order.

Steps 380 may be utilized to calibrate the modulator phase and may comprise the steps 304 through 316. In step 304, the in-phase baseband signal s_(I)(t) may be set to zero, as described for FIG. 2A. In step 306, the quadrature baseband signal s_(Q)(t) may be, for example, a constant power pilot tone that may be suitably chosen for power detection in a I-channel RSSI block, for example I-channel RSSI block 214. In step 308, the TX programmable phase shift, for example 202, may be initialized to a starting phase value. For example, this may be a minimum phase setting. In step 310, the signal power for the initial setting of the TX phase shift may be measured in the I-channel RSSI block 214, for example, as described in FIG. 2. The phase setting and the resulting signal power measured by the I-channel RSSI block 214 may be recorded for example in the device control 206. In step 312, the phase may be adjusted to a new value. This new value may be a slight offset with respect to the previous value. In step 314, if the RSSI measurements have not been taken over the entire desired phase range, the RSSI measurements may continue at various phase shifts. If the RSSI measurements have been taken over the entire desired phase range, the TX programmable phase shift 202 may be set to a desirable phase value that may lead to, for example, a minimum RSSI power as measured by the I-channel RSSI block 214.

Steps 350 to calibrate the demodulator phase may comprise the steps 318 through 330. In step 318, the quadrature baseband signal S_(Q)(t) may be set to zero, as described for FIG. 2A. In step 320, the in-phase baseband signal s_(I)(t) may be, for example, a constant power pilot tone that may be suitably chosen for power detection in a Q-channel RSSI block, for example Q-channel RSSI block 210. In step 322, the RX programmable phase shift, for example 204, may be initialized to a starting phase value. For example, this may be a minimum phase setting. In step 324, the signal power for the initial setting of the RX phase shift may be measured in the Q-channel RSSI block 210, for example, as described in FIG. 2A. The phase setting and the resulting signal power measured by the Q-channel RSSI block 210 may be recorded, for example in the device control 206. In step 326, the phase may be adjusted to a new value. This new value may be a slight offset with respect to the previous value. In step 328, if the RSSI measurements have not been taken over the entire desired phase range, the RSSI measurements may continue at various phase shifts. If the RSSI measurements have been taken over the entire desired phase range, the RX programmable phase shift 204 may be set to a desirable phase value that may lead to, for example, a minimum RSSI power as measured by the Q-channel RSSI block 210.

In accordance with an embodiment of the invention, a method and system for calibration in an FM transceiver system 100 may comprise: in an integrated FM system 100 comprising an FM radio transmitter 180, an FM radio receiver 150 and a common local oscillator 108, generating via the common local oscillator 108, one or more RF carrier signals, for example cos(w_(c)t), and corresponding phase-shifted versions of the generated one or more RF carrier signals, for example −sin(w_(c)t+e_(t)). The FM radio transmitter 180 and/or the FM radio receiver 150 may be calibrated based on an RF calibration signal s(t) generated from the one or more RF carrier signals and/or the corresponding phase-shifted versions of the generated one or more RF carrier signals, as described for FIGS. 2A and 2B.

A phase between the one or more RF carrier signals, for example cos(w_(c)t) and the corresponding phase-shifted versions of the generated one or more RF carrier signals, for example −sin(w_(c)t+e_(t)), may be adjusted based on the generated RF calibration signal s(t), for example in the TX programmable phase shift 202. An in-phase baseband signal component s_(I)(t) associated with the generated RF calibration signal s(t) may be zeroed. A signal power of an in-phase signal component after demodulating the generated RF calibration signal s(t) may be measured, for example r_(I)(t). A quadrature baseband signal component, for example S_(Q)(t), associated with the generated RF calibration signal s(t) may be zeroed. A signal power of a quadrature signal component after demodulating the generated RF calibration signal s(t) may be measured, for example r_(Q)(t). Based on the generated RF calibration signal s(t), a tuning range and/or gain stages of the FM radio receiver and/or the FM radio transmitter may be adjusted and/or calibrated.

The FM radio transmitter 180 and/or the FM radio receiver 150 may be calibrated based on the calibration RF signal s(t). A phase between the one or more RF carrier signals, for example cos(w_(c)t) and the phase-shifted versions thereof, for example −sin(w_(c)t+e_(t)), may be adjusted, for example in the TX programmable phase shift 202, based on at least the calibration RF signal s(t). An in-phase baseband signal component s_(I)(t) associated with the calibration RF signal s(t) may be zeroed to achieve the phase adjusting. A signal power of an in-phase signal component r_(I)(t) after demodulating the calibration RF signal s(t) may be measured to achieve the phase adjusting. A quadrature baseband signal component S_(Q)(t) associated with the calibration RF signal s(t) may be zeroed to achieve the phase adjusting. A signal power of a quadrature signal component r_(Q)(t) resulting from demodulation of the calibration RF signal s(t) may be measured to achieve the phase adjusting. Based on the calibration RF signal s(t), a tuning range and/or gain stages of the FM radio receiver may be adjusted and/or calibrated.

Another embodiment of the invention may provide a machine-readable storage, having stored thereon, a computer program having at least one code section executable by a machine, thereby causing the machine to perform the steps as described herein for calibration in an FM transceiver system.

Accordingly, the present invention may be realized in hardware, software, or a combination of hardware and software. The present invention may be realized in a centralized fashion in at least one computer system, or in a distributed fashion where different elements are spread across several interconnected computer systems. Any kind of computer system or other apparatus adapted for carrying out the methods described herein is suited. A typical combination of hardware and software may be a general-purpose computer system with a computer program that, when being loaded and executed, controls the computer system such that it carries out the methods described herein.

The present invention may also be embedded in a computer program product, which comprises all the features enabling the implementation of the methods described herein, and which when loaded in a computer system is able to carry out these methods. Computer program in the present context means any expression, in any language, code or notation, of a set of instructions intended to cause a system having an information processing capability to perform a particular function either directly or after either or both of the following: a) conversion to another language, code or notation; b) reproduction in a different material form.

While the present invention has been described with reference to certain embodiments, it will be understood by those skilled in the art that various changes may be made and equivalents may be substituted without departing from the scope of the present invention. In addition, many modifications may be made to adapt a particular situation or material to the teachings of the present invention without departing from its scope. Therefore, it is intended that the present invention not be limited to the particular embodiment disclosed, but that the present invention will include all embodiments falling within the scope of the appended claims. 

1. A method for processing communication signals, the method comprising: in an integrated FM system comprising an FM radio transmitter, an FM radio receiver and a common local oscillator, generating via said common local oscillator, one or more RF carrier signals and corresponding phase-shifted versions of said generated one or more RF carrier signals; and calibrating said FM radio transmitter and/or said FM radio receiver based on an RF calibration signal generated from said one or more RF carrier signals and/or said corresponding phase-shifted versions of said generated one or more RF carrier signals.
 2. The method according to claim 1, comprising adjusting a phase between said one or more RF carrier signals and said corresponding phase-shifted versions of said generated one or more RF carrier signals, based on said generated RF calibration signal.
 3. The method according to claim 2, comprising zeroing an in-phase baseband signal component associated with said generated RF calibration signal.
 4. The method according to claim 3, comprising measuring a signal power of an in-phase signal component after demodulating said generated RF calibration signal.
 5. The method according to claim 2, comprising zeroing a quadrature baseband signal component associated with said generated RF calibration signal.
 6. The method according to claim 5, comprising measuring a signal power of an in-phase signal component after demodulating said generated RF calibration signal.
 7. The method according to claim 1, comprising adjusting a tuning range of said FM radio receiver based on said generated RF calibration signal
 8. The method according to claim 1, comprising calibrating a tuning range of said FM radio transmitter based on said generated RF calibration signal.
 9. The method according to claim 1, comprising adjusting one or more gain stages in said FM radio receiver based on said generated RF calibration signal.
 10. The method according to claim 1, comprising calibrating one or more gain stages in said FM radio transmitter based on said generated RF calibration signal.
 11. The method according to claim 1, comprising calibrating one or more gain stages in said FM radio receiver based on said generated RF calibration signal.
 12. A system for processing communication signals, the system comprising: one or more circuits in an integrated FM system comprising an FM radio transmitter, an FM radio receiver and a common local oscillator, said one or more circuits enable: generating via said common local oscillator, one or more RF carrier signals and corresponding phase-shifted versions of said generated one or more RF carrier signals; and calibration of said FM radio transmitter and/or said FM radio receiver based on an RF calibration signal generated from said one or more RF carrier signals and/or said corresponding phase-shifted versions of said generated one or more RF carrier signals.
 13. The system according to claim 12, wherein said one or more circuits adjust a phase between said one or more RF carrier signals and said corresponding phase-shifted versions of said generated one or more RF carrier signals, based on said generated RF calibration signal.
 14. The system according to claim 13, wherein said one or more circuits zero an in-phase baseband signal component associated with said generated RF calibration signal.
 15. The system according to claim 14, wherein said one or more circuits measure a signal power of an in-phase signal component after demodulating said generated RF calibration signal.
 16. The system according to claim 13, wherein said one or more circuits zero a quadrature baseband signal component associated with said generated RF calibration signal.
 17. The system according to claim 16, wherein said one or more circuits measure a signal power of an in-phase signal component after demodulating said generated RF calibration signal.
 18. The system according to claim 12, wherein said one or more circuits adjust a tuning range of said FM radio receiver based on said generated RF calibration signal
 19. The system according to claim 12, wherein said one or more circuits calibrate a tuning range of said FM radio receiver based on said generated RF calibration signal.
 20. The system according to claim 12, wherein said one or more circuits adjust one or more gain stages in said FM radio receiver based on said generated RF calibration signal.
 21. The system according to claim 12, wherein said one or more circuits calibrate one or more gain stages in said FM transmitter based on said generated RF calibration signal.
 22. The method according to claim 12, wherein said one or more circuits calibrate one or more gain stages in said FM radio receiver based on said generated RF calibration signal. 